CA2202457A1 - Microwave phase shifter including a reflective phase shift stage and a frequency multiplication stage - Google Patents
Microwave phase shifter including a reflective phase shift stage and a frequency multiplication stageInfo
- Publication number
- CA2202457A1 CA2202457A1 CA002202457A CA2202457A CA2202457A1 CA 2202457 A1 CA2202457 A1 CA 2202457A1 CA 002202457 A CA002202457 A CA 002202457A CA 2202457 A CA2202457 A CA 2202457A CA 2202457 A1 CA2202457 A1 CA 2202457A1
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- 230000010363 phase shift Effects 0.000 title claims abstract description 20
- 230000005669 field effect Effects 0.000 claims abstract description 6
- 238000000034 method Methods 0.000 claims description 8
- 238000002347 injection Methods 0.000 claims description 3
- 239000007924 injection Substances 0.000 claims description 3
- JBRZTFJDHDCESZ-UHFFFAOYSA-N AsGa Chemical compound [As]#[Ga] JBRZTFJDHDCESZ-UHFFFAOYSA-N 0.000 description 5
- 229910001218 Gallium arsenide Inorganic materials 0.000 description 4
- 230000003071 parasitic effect Effects 0.000 description 4
- 238000010586 diagram Methods 0.000 description 3
- 230000000694 effects Effects 0.000 description 2
- PNEYBMLMFCGWSK-UHFFFAOYSA-N aluminium oxide Inorganic materials [O-2].[O-2].[O-2].[Al+3].[Al+3] PNEYBMLMFCGWSK-UHFFFAOYSA-N 0.000 description 1
- 238000003491 array Methods 0.000 description 1
- 239000003990 capacitor Substances 0.000 description 1
- 238000006243 chemical reaction Methods 0.000 description 1
- 230000002401 inhibitory effect Effects 0.000 description 1
- 229910052751 metal Inorganic materials 0.000 description 1
- 239000002184 metal Substances 0.000 description 1
- 239000000758 substrate Substances 0.000 description 1
Classifications
-
- H—ELECTRICITY
- H01—ELECTRIC ELEMENTS
- H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
- H01P1/00—Auxiliary devices
- H01P1/18—Phase-shifters
- H01P1/185—Phase-shifters using a diode or a gas filled discharge tube
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
- H03H11/00—Networks using active elements
- H03H11/02—Multiple-port networks
- H03H11/16—Networks for phase shifting
- H03H11/20—Two-port phase shifters providing an adjustable phase shift
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- Digital Transmission Methods That Use Modulated Carrier Waves (AREA)
- Networks Using Active Elements (AREA)
Abstract
A phase shifter/modulator which provides linear phase control of a microwave continuous wave carrier signal.
Linear phase shift range in excess of 360 degrees is provided as a result of linear variation in control voltage. A reference signal, at a subharmonic of the output carrier signal frequency, is injected into a highly linear fractional range reflection type phase shifter.
This fractional phase modulated signal is input to a field effect transistor (FET) based frequency/phase multiplier.
The frequency/phase multiplier translates the subharmonic reference signal to the desired output carrier frequency with the full 360 degree phase shift range imposed on the output carrier signal.
Linear phase shift range in excess of 360 degrees is provided as a result of linear variation in control voltage. A reference signal, at a subharmonic of the output carrier signal frequency, is injected into a highly linear fractional range reflection type phase shifter.
This fractional phase modulated signal is input to a field effect transistor (FET) based frequency/phase multiplier.
The frequency/phase multiplier translates the subharmonic reference signal to the desired output carrier frequency with the full 360 degree phase shift range imposed on the output carrier signal.
Description
CA 022024~7 1997-04-11 TITLE OF THE INVENTION:
Microwave Phase Shifter NAMES OF INVENTORS:
David M. Klymyshyn Surinder Kumar Abbas Mohammadi FIELD OF THE INVENTION
This invention relates to phase shifters, particularly for use in microwave circuit applications.
BACKGROUND OF THE INVENTION
This invention finds use in a number of microwave circuit applications which require full 360 degree linear phase control of a carrier signal. Examples of such applications include continuous phase modulation (CPM) and indirect frequency modulation (FM) radio communication transmitters, phase synchronisation of antenna and oscillator arrays, and phased array antenna beam steering.
Most microwave linear phase shifters are based on a single stage reflection topology using a circulator or coupler with appropriate reflective terminations. Large range phase shifters at these frequencies have been designed with some success as a cascade of smaller range single stage shifters. Another method of obtaining large range linear phase shift at microwave frequency is by detuning the tank circuit of an injection locked oscillator.
The problem with these traditional implementation strategies is that the circuitry required to implement the full 360 degree phase shift range becomes quite complicated. At microwave frequencies, this complicated circuitry can become quite inhibitive and can reduce the CA 022024~7 1997-04-11 performance significantly on implementation. Also, phase shift linearity has to be sacrificed in order to maintain acceptable bandwidth. As a result of poor phase shift linearity over the full 360 degree range, complicated predistortion circuitry is required on the phase control signal to realize acceptable phase shift linearity.
SUMMARY OF THE INVENTION
This invention presents an alternative approach to realizing full 360 degree microwave linear phase shifters.
A subharmonic reference signal, at 1/N times the carrier frequency, is injected into a highly linear fractional range (360/N degree) reflection type varactor phase shifter. This fractional CPM carrier is then fed to a field effect transistor (FET) based linear frequency/phase multiplier (xN). The frequency/phase multiplier translates r the subharmonic reference frequency to the desired carrier frequency and restores the full 360 degree phase shift range. The benefits of this new implementation include:
a. Simplified, cost effective hardware architecture, requiring only a single stage reflection type phase shifter.
b. Use of a subharmonic reference signal, which is easier to obtain than the higher frequency carrier signal.
c. Implementation of the linear reflection phase shifter at a subharmonic of the carrier frequency, resulting in realistic varactor values for the reactive terminations and higher phase shift linearity.
d. An effective xN increase in output bandwidth from the frequency/phase multiplier, when compared with a phase shifter designed at the carrier frequency.
There is therefore provided in accordance with one aspect of the invention, a phase shifting device comprising a voltage controlled phase shifter having an input port for CA 022024~7 1997-04-11 injection of a first signal having a first frequency and a first frequency multiplier operatively connected to the voltage controlled phase shifter for receiving the first signal from the voltage controlled phase shifter and for translating the first signal to a second signal having a second frequency, the second frequency being higher than the first frequency.
In a further aspect of the invention, there is provided a method of phase shifting a signal comprising the steps of phase shifting a first signal at a first frequency and translating the first frequency to a second frequency, higher than the first frequency.
In further aspects of the invention: the phase shifter preferably operates at microwave frequencies, the multiplier is formed with a field effect transistor, the multiplier has an integral multiplication factor; the voltage controlled phase shifter comprises a quadrature coupler having a pair of reflection ports and each of the reflection ports is terminated by equal reactive terminations; the reactive terminations comprise reverse biased varactor diodes, preferably abrupt or hyperabrupt varactor diodes with grounded series inductors; to achieve higher multiplication factors, a second frequency multiplier may be operatively connected to the first frequency multiplier for receiving the second signal and for translating the second signal to a third signal having a third frequency, the third frequency being higher than the second frequency, the third signal preferably being a multiple of the second frequency.
In a still further aspect of the invention, the first frequency is translated in a FET, having a gate bias, the first signal has a signal level, and the gate bias and subharmonic input signal level are selected such that the CA 022024~7 1997-04-11 FET has unconditional stability at all subharmonics of the second signal.
These and other aspects of the invention are described in the detailed description of the invention and claimed in the claims that follow.
BRIEF DESCRIPTION OF THE DRAWINGS
There will now be described preferred embodiments of the invention, with reference to the drawings, by way of illustration only and not with the intention of limiting the scope of the invention, in which like numerals denote like elements and in which:
Fig. 1 is a functional block diagram of the 360 degree linear phase shifter;
Fig. 2 is a microstrip schematic for a 360 degree linear phase shifter; and Fig. 3 is a functional block diagram showing an implementation of the invention with multiple multiplier stages.
DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS
The present invention is a novel device for achieving full 360 degree linear phase control of a microwave carrier signal or other microwave signal. In this patent document, microwave is taken to mean the range of frequencies from 1-100 GHz although it will be appreciated that the utility of the invention in higher frequencies is only limited by the present availability of components, and not by the principle of operation of the invention.
The functional block diagram of the 360 degree linear phase shifter is shown in Figure 1. Figure 2 presents a microstrip schematic for a 360 degree linear phase shifter implementation example at 18 GHz, using gallium arsenide (GaAs) hyperabrupt junction varactors and CA 022024~7 1997-04-11 a single GaAs FET. Very few components are required to implement this invention at microwave frequencies, which makes it attractive for many applications.
The phase shifting device in one embodiment of the invention has two main functional elements. The first is a voltage controlled phase shifter 10 to which is operatively connected the second element which is a frequency multiplier 12 for receiving phase shifted signals from the voltage controlled phase shifter 10 and translating the phase shifted signals to a higher frequency, thus expanding the effective phase shift.
Referring to the figures, a subharmonic reference signal from source 11, at 1/5 of the carrier frequency or 3.6 GHz for the sample implementation, is injected into the voltage controlled phase shifter 10 at -12 dBm. The phase shifter 10 is preferably a linear fractional range reflection type varactor phase shifter. The phase shifter 10 is comprised of a microstrip quadrature hybrid coupler 14, with a direct port 16 and coupled port 18 terminated in the same variable capacitance series inductance, inductor/capacitor (LC), reactive terminations 36, 38.
The quadrature hybrid 14 is not especially wideband, but is simple, and provides adequate bandwidth for the sample implementation. Other wideband couplers or circulators could also be used. The variable capacitance characteristic is provided via reverse biased varactor diodes 20, 21.
With ideal reactive terminations, all power is reflected from the direct and coupled ports 16, 18 and combines constructively at the isolated port 19 of the coupler 14 with no loss. In practice, some loss occurs from parasitic resistance in the terminations. The phase shift through the hybrid coupler 14 is a function of the termination reactance, and thus, can be controlled by varying the bias CA 022024~7 1997-04-11 voltage on the varactors 20, 21 and changing the series capacitance.
The reference signal enters the input port 24 of the microstrip quadrature hybrid coupler 14 and is reflected to the isolated output port 19 of the coupler 14 by equivalent reactive terminations 36, 38 on the direct and coupled ports 16, 18 of the coupler 14. The reactive terminations are comprised of grounded series inductors 40, 42 and series reverse biased varactor diodes 20, 21. The varactor diodes 20, 21 provide a variable capacitance characteristic by varying the reverse biased diode control voltage supplied at 44 through inductor 45 (Fig. 2). The total series inductance includes the parasitic series inductance of the varactor diodes 20, 21 and ground connections.
The varactor capacitance versus voltage (CV) relationship that provides linear phase shift through the hybrid 14 is not linear. Obtaining linear phase shift from a reflective termination requires that the termination reactance be matched to the tangent function of the linear varactor bias voltage variation. An abrupt junction varactor, with gamma (~) of 0.5, is generally suitable for matching to the tangent function, over a limited bias voltage range. With a subharmonic reference signal in the range of 3 to 5 GHz, the required minimum varactor capacitance becomes comparable to the varactor parasitic package capacitance. The result of this is a flattening of the CV characteristic and the termination reactance characteristic as a function of increasing bias voltage.
With this si~uation, the abrupt junction varactor, with g=
0.5, no lon~er provides a good match to the tangent function, resulting in poor phase shift linearity. Using an available hyperabrupt junction varactor for the diodes 20, 21, with g= 0.75, a near optimal match to the tangent function is obtained over a limited phase shift range, when CA 022024~7 1997-04-11 a suitable series inductor 40, 42 is used, for a subharmonic reference signal of 3.6 GHz.
The fractional CPM signal at the output 22 of the reflection phase shifter 10 is fed to a field effect transistor (FET) frequency/phase multiplier 12. The frequency/phase multiplier 12 translates the modulated subharmonic reference signal to the desired carrier frequency and restores the full 360 degree phase shift range. The multiplier 12 consists of a GaAs FET 26, input and output matching circuitry 28, 30, output harmonic termination circuitry 32, and biasing circuitry (not shown, but various implementations may be used as is known to a skilled person in the art). The FET 26 is biased strongly Class C to obtain a conduction angle at the gate that m~x;m; zes the 5th harmonic. Input and output matching circuitry 28, 30 should be designed to provide simultaneous conjugate match at 3.6 GHz on the input and 18 GHz on the output. A simple coupled line bandpass filter (BPF) 32 is provided on the output to reject all unwanted spurious harmonic signals. More complicated harmonic termination circuitry on the FET input and output could also be used but was unnecessary for the sample implementation.
The input matching circuitry 28 is preferably implemented using microstrip as shown in Fig. 2, and is provided to conjugately match the phase shifter output impedance to the low input gate impedance of the FET 26 at the subharmonic reference frequency. The input impedance of the FET 26 is essentially reactive, which makes wideband input matching very difficult. The effect of mismatch at the FET gate 46 is a variation in gate signal level. This, together with non-constant phase shifter output impedance as a function of control voltage causes a variation in the FET conduction angle. The harmonic output level in high harmonic FET multipliers is fairly sensitive to conduction CA 022024~7 1997-04-11 angle, so significant amplitude modulation (AM) can be a result in the output CPM signal if this input matching is poor. This effect is offset by the increase in effective output bandwidth by a factor of xN as a result of frequency multiplication, so a narrowband subharmonic gate match is generally acceptable.
The multiplication factor for the sample implementation is x5. If multiplication factors greater than x7 are required, an additional frequency multiplier 50, constructed as the multiplier 12 with a Class C biased FET stage 26, along with interstage matching and harmonic termination circuitry, can be added as shown in Fig. 3.
The gate 46 of the FET 26 is biased for a realistic conduction angle that maximizes the 5th harmonic output level (about 140 degrees). The gate bias and the FET input signal level should be carefully selected according to known techniques to provide unconditional stability at all subharmonics of the output carrier frequency. If the combination of gate bias and input signal amplitude pulls the peak gate voltage too far below the gate threshold voltage, the transistor becomes conditionally stable at subharmonic frequencies of the output, and is very difficult to terminate. The input and output microstrip matching networks 28 and 30 should be designed to provide simultaneous conjugate matching to the FET at 3.6 GHz on the input and 18 GHz on the output in the exemplary implementation. A 2nd order coupled line bandpass filter (BPF) 32 is used on the output of the matching network 30 to select the desired CPM carrier signal and reject all unwanted spurious harmonic signals. The filter rejection is adequate to maintain all spurious harmonic output signals at below -30 dBc. The CPM carrier signal 48 with 360 degree linear phase modulation range is output from the FET multiplier 12 at a level of approximately -30 dBm.
CA 022024~7 1997-04-11 Using this novel hardware architecture, a sample circuit was designed at 18 GHz using microstrip as is shown for example at 34 in Fig. 2 for a sample 360 degree linear phase shifter implementation at 18 GHz, on a 25 mil thick Alumina substrate (not shown) with ~r=9.8 and metal thickness of 0.15 mil. Hyperabrupt, GaAs chip varactors 20, 21, with ~= 0.75, Cmax= 2.4 pF, and parasitic package capacitance of Cp= 0.15 pF, were used in the fractional phase shifter reactive terminations 36, 38 along with 3.5 nH of series inductance. A general purpose medium power GaAs FET 26 was used in the frequency/phase multiplier section.
The circuit provided a total phase shift range of 425 degrees, for a varactor reverse bias voltage range of 1.3 to 11.8 V. Residual amplitude modulation (AM) is within 0.15 dB over the same bias range. A linear phase shift range of 370 degrees at 18 GHz was obtained within +1 degree, for a varactor reverse bias voltage range of 1.4 to 10.5 V. The output bandwidth at 18 GHz is 200 MHz, with amplitude and phase distortion within 0.5 dB and 5 degrees, respectively, across the band. All output harmonic spurious signals are below -30 dBc, and the circuit conversion loss is approximately 18 dB.
While a preferred implementation has been described, the invention is not limited to the exemplary features described. A person skilled in the art will appreciate that immaterial variations are intended to be encompassed within the scope of the invention.
Microwave Phase Shifter NAMES OF INVENTORS:
David M. Klymyshyn Surinder Kumar Abbas Mohammadi FIELD OF THE INVENTION
This invention relates to phase shifters, particularly for use in microwave circuit applications.
BACKGROUND OF THE INVENTION
This invention finds use in a number of microwave circuit applications which require full 360 degree linear phase control of a carrier signal. Examples of such applications include continuous phase modulation (CPM) and indirect frequency modulation (FM) radio communication transmitters, phase synchronisation of antenna and oscillator arrays, and phased array antenna beam steering.
Most microwave linear phase shifters are based on a single stage reflection topology using a circulator or coupler with appropriate reflective terminations. Large range phase shifters at these frequencies have been designed with some success as a cascade of smaller range single stage shifters. Another method of obtaining large range linear phase shift at microwave frequency is by detuning the tank circuit of an injection locked oscillator.
The problem with these traditional implementation strategies is that the circuitry required to implement the full 360 degree phase shift range becomes quite complicated. At microwave frequencies, this complicated circuitry can become quite inhibitive and can reduce the CA 022024~7 1997-04-11 performance significantly on implementation. Also, phase shift linearity has to be sacrificed in order to maintain acceptable bandwidth. As a result of poor phase shift linearity over the full 360 degree range, complicated predistortion circuitry is required on the phase control signal to realize acceptable phase shift linearity.
SUMMARY OF THE INVENTION
This invention presents an alternative approach to realizing full 360 degree microwave linear phase shifters.
A subharmonic reference signal, at 1/N times the carrier frequency, is injected into a highly linear fractional range (360/N degree) reflection type varactor phase shifter. This fractional CPM carrier is then fed to a field effect transistor (FET) based linear frequency/phase multiplier (xN). The frequency/phase multiplier translates r the subharmonic reference frequency to the desired carrier frequency and restores the full 360 degree phase shift range. The benefits of this new implementation include:
a. Simplified, cost effective hardware architecture, requiring only a single stage reflection type phase shifter.
b. Use of a subharmonic reference signal, which is easier to obtain than the higher frequency carrier signal.
c. Implementation of the linear reflection phase shifter at a subharmonic of the carrier frequency, resulting in realistic varactor values for the reactive terminations and higher phase shift linearity.
d. An effective xN increase in output bandwidth from the frequency/phase multiplier, when compared with a phase shifter designed at the carrier frequency.
There is therefore provided in accordance with one aspect of the invention, a phase shifting device comprising a voltage controlled phase shifter having an input port for CA 022024~7 1997-04-11 injection of a first signal having a first frequency and a first frequency multiplier operatively connected to the voltage controlled phase shifter for receiving the first signal from the voltage controlled phase shifter and for translating the first signal to a second signal having a second frequency, the second frequency being higher than the first frequency.
In a further aspect of the invention, there is provided a method of phase shifting a signal comprising the steps of phase shifting a first signal at a first frequency and translating the first frequency to a second frequency, higher than the first frequency.
In further aspects of the invention: the phase shifter preferably operates at microwave frequencies, the multiplier is formed with a field effect transistor, the multiplier has an integral multiplication factor; the voltage controlled phase shifter comprises a quadrature coupler having a pair of reflection ports and each of the reflection ports is terminated by equal reactive terminations; the reactive terminations comprise reverse biased varactor diodes, preferably abrupt or hyperabrupt varactor diodes with grounded series inductors; to achieve higher multiplication factors, a second frequency multiplier may be operatively connected to the first frequency multiplier for receiving the second signal and for translating the second signal to a third signal having a third frequency, the third frequency being higher than the second frequency, the third signal preferably being a multiple of the second frequency.
In a still further aspect of the invention, the first frequency is translated in a FET, having a gate bias, the first signal has a signal level, and the gate bias and subharmonic input signal level are selected such that the CA 022024~7 1997-04-11 FET has unconditional stability at all subharmonics of the second signal.
These and other aspects of the invention are described in the detailed description of the invention and claimed in the claims that follow.
BRIEF DESCRIPTION OF THE DRAWINGS
There will now be described preferred embodiments of the invention, with reference to the drawings, by way of illustration only and not with the intention of limiting the scope of the invention, in which like numerals denote like elements and in which:
Fig. 1 is a functional block diagram of the 360 degree linear phase shifter;
Fig. 2 is a microstrip schematic for a 360 degree linear phase shifter; and Fig. 3 is a functional block diagram showing an implementation of the invention with multiple multiplier stages.
DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS
The present invention is a novel device for achieving full 360 degree linear phase control of a microwave carrier signal or other microwave signal. In this patent document, microwave is taken to mean the range of frequencies from 1-100 GHz although it will be appreciated that the utility of the invention in higher frequencies is only limited by the present availability of components, and not by the principle of operation of the invention.
The functional block diagram of the 360 degree linear phase shifter is shown in Figure 1. Figure 2 presents a microstrip schematic for a 360 degree linear phase shifter implementation example at 18 GHz, using gallium arsenide (GaAs) hyperabrupt junction varactors and CA 022024~7 1997-04-11 a single GaAs FET. Very few components are required to implement this invention at microwave frequencies, which makes it attractive for many applications.
The phase shifting device in one embodiment of the invention has two main functional elements. The first is a voltage controlled phase shifter 10 to which is operatively connected the second element which is a frequency multiplier 12 for receiving phase shifted signals from the voltage controlled phase shifter 10 and translating the phase shifted signals to a higher frequency, thus expanding the effective phase shift.
Referring to the figures, a subharmonic reference signal from source 11, at 1/5 of the carrier frequency or 3.6 GHz for the sample implementation, is injected into the voltage controlled phase shifter 10 at -12 dBm. The phase shifter 10 is preferably a linear fractional range reflection type varactor phase shifter. The phase shifter 10 is comprised of a microstrip quadrature hybrid coupler 14, with a direct port 16 and coupled port 18 terminated in the same variable capacitance series inductance, inductor/capacitor (LC), reactive terminations 36, 38.
The quadrature hybrid 14 is not especially wideband, but is simple, and provides adequate bandwidth for the sample implementation. Other wideband couplers or circulators could also be used. The variable capacitance characteristic is provided via reverse biased varactor diodes 20, 21.
With ideal reactive terminations, all power is reflected from the direct and coupled ports 16, 18 and combines constructively at the isolated port 19 of the coupler 14 with no loss. In practice, some loss occurs from parasitic resistance in the terminations. The phase shift through the hybrid coupler 14 is a function of the termination reactance, and thus, can be controlled by varying the bias CA 022024~7 1997-04-11 voltage on the varactors 20, 21 and changing the series capacitance.
The reference signal enters the input port 24 of the microstrip quadrature hybrid coupler 14 and is reflected to the isolated output port 19 of the coupler 14 by equivalent reactive terminations 36, 38 on the direct and coupled ports 16, 18 of the coupler 14. The reactive terminations are comprised of grounded series inductors 40, 42 and series reverse biased varactor diodes 20, 21. The varactor diodes 20, 21 provide a variable capacitance characteristic by varying the reverse biased diode control voltage supplied at 44 through inductor 45 (Fig. 2). The total series inductance includes the parasitic series inductance of the varactor diodes 20, 21 and ground connections.
The varactor capacitance versus voltage (CV) relationship that provides linear phase shift through the hybrid 14 is not linear. Obtaining linear phase shift from a reflective termination requires that the termination reactance be matched to the tangent function of the linear varactor bias voltage variation. An abrupt junction varactor, with gamma (~) of 0.5, is generally suitable for matching to the tangent function, over a limited bias voltage range. With a subharmonic reference signal in the range of 3 to 5 GHz, the required minimum varactor capacitance becomes comparable to the varactor parasitic package capacitance. The result of this is a flattening of the CV characteristic and the termination reactance characteristic as a function of increasing bias voltage.
With this si~uation, the abrupt junction varactor, with g=
0.5, no lon~er provides a good match to the tangent function, resulting in poor phase shift linearity. Using an available hyperabrupt junction varactor for the diodes 20, 21, with g= 0.75, a near optimal match to the tangent function is obtained over a limited phase shift range, when CA 022024~7 1997-04-11 a suitable series inductor 40, 42 is used, for a subharmonic reference signal of 3.6 GHz.
The fractional CPM signal at the output 22 of the reflection phase shifter 10 is fed to a field effect transistor (FET) frequency/phase multiplier 12. The frequency/phase multiplier 12 translates the modulated subharmonic reference signal to the desired carrier frequency and restores the full 360 degree phase shift range. The multiplier 12 consists of a GaAs FET 26, input and output matching circuitry 28, 30, output harmonic termination circuitry 32, and biasing circuitry (not shown, but various implementations may be used as is known to a skilled person in the art). The FET 26 is biased strongly Class C to obtain a conduction angle at the gate that m~x;m; zes the 5th harmonic. Input and output matching circuitry 28, 30 should be designed to provide simultaneous conjugate match at 3.6 GHz on the input and 18 GHz on the output. A simple coupled line bandpass filter (BPF) 32 is provided on the output to reject all unwanted spurious harmonic signals. More complicated harmonic termination circuitry on the FET input and output could also be used but was unnecessary for the sample implementation.
The input matching circuitry 28 is preferably implemented using microstrip as shown in Fig. 2, and is provided to conjugately match the phase shifter output impedance to the low input gate impedance of the FET 26 at the subharmonic reference frequency. The input impedance of the FET 26 is essentially reactive, which makes wideband input matching very difficult. The effect of mismatch at the FET gate 46 is a variation in gate signal level. This, together with non-constant phase shifter output impedance as a function of control voltage causes a variation in the FET conduction angle. The harmonic output level in high harmonic FET multipliers is fairly sensitive to conduction CA 022024~7 1997-04-11 angle, so significant amplitude modulation (AM) can be a result in the output CPM signal if this input matching is poor. This effect is offset by the increase in effective output bandwidth by a factor of xN as a result of frequency multiplication, so a narrowband subharmonic gate match is generally acceptable.
The multiplication factor for the sample implementation is x5. If multiplication factors greater than x7 are required, an additional frequency multiplier 50, constructed as the multiplier 12 with a Class C biased FET stage 26, along with interstage matching and harmonic termination circuitry, can be added as shown in Fig. 3.
The gate 46 of the FET 26 is biased for a realistic conduction angle that maximizes the 5th harmonic output level (about 140 degrees). The gate bias and the FET input signal level should be carefully selected according to known techniques to provide unconditional stability at all subharmonics of the output carrier frequency. If the combination of gate bias and input signal amplitude pulls the peak gate voltage too far below the gate threshold voltage, the transistor becomes conditionally stable at subharmonic frequencies of the output, and is very difficult to terminate. The input and output microstrip matching networks 28 and 30 should be designed to provide simultaneous conjugate matching to the FET at 3.6 GHz on the input and 18 GHz on the output in the exemplary implementation. A 2nd order coupled line bandpass filter (BPF) 32 is used on the output of the matching network 30 to select the desired CPM carrier signal and reject all unwanted spurious harmonic signals. The filter rejection is adequate to maintain all spurious harmonic output signals at below -30 dBc. The CPM carrier signal 48 with 360 degree linear phase modulation range is output from the FET multiplier 12 at a level of approximately -30 dBm.
CA 022024~7 1997-04-11 Using this novel hardware architecture, a sample circuit was designed at 18 GHz using microstrip as is shown for example at 34 in Fig. 2 for a sample 360 degree linear phase shifter implementation at 18 GHz, on a 25 mil thick Alumina substrate (not shown) with ~r=9.8 and metal thickness of 0.15 mil. Hyperabrupt, GaAs chip varactors 20, 21, with ~= 0.75, Cmax= 2.4 pF, and parasitic package capacitance of Cp= 0.15 pF, were used in the fractional phase shifter reactive terminations 36, 38 along with 3.5 nH of series inductance. A general purpose medium power GaAs FET 26 was used in the frequency/phase multiplier section.
The circuit provided a total phase shift range of 425 degrees, for a varactor reverse bias voltage range of 1.3 to 11.8 V. Residual amplitude modulation (AM) is within 0.15 dB over the same bias range. A linear phase shift range of 370 degrees at 18 GHz was obtained within +1 degree, for a varactor reverse bias voltage range of 1.4 to 10.5 V. The output bandwidth at 18 GHz is 200 MHz, with amplitude and phase distortion within 0.5 dB and 5 degrees, respectively, across the band. All output harmonic spurious signals are below -30 dBc, and the circuit conversion loss is approximately 18 dB.
While a preferred implementation has been described, the invention is not limited to the exemplary features described. A person skilled in the art will appreciate that immaterial variations are intended to be encompassed within the scope of the invention.
Claims (22)
1. A phase shifting device, comprising:
a voltage controlled phase shifter having an input port for injection of a first signal having a first frequency, the voltage controlled phase shifter having an output port; and a first frequency multiplier operatively connected to the voltage controlled phase shifter for receiving the first signal from the output port of the voltage controlled phase shifter and for translating the first signal to a second signal having a second frequency, the second frequency being higher than the first frequency.
a voltage controlled phase shifter having an input port for injection of a first signal having a first frequency, the voltage controlled phase shifter having an output port; and a first frequency multiplier operatively connected to the voltage controlled phase shifter for receiving the first signal from the output port of the voltage controlled phase shifter and for translating the first signal to a second signal having a second frequency, the second frequency being higher than the first frequency.
2. The phase shifting device of claim 1 in which the source of the first signal is a source of a signal having a frequency in the range from 1 GHz to 100 GHz.
3. The phase shifting device of claim 2 in which the first frequency multiplier has an integral multiplication factor.
4. The phase shifting device of claim 3 in which the voltage controlled phase shifter is a linear phase shifter.
5. The phase shifting device of claim 4 in which the voltage controlled phase shifter comprises:
a quadrature coupler having a pair of reflection ports; and each of the reflection ports being terminated by equal reactive terminations.
a quadrature coupler having a pair of reflection ports; and each of the reflection ports being terminated by equal reactive terminations.
6. The phase shifting device of claim 5 in which the reactive terminations comprise reverse biased varactor diodes.
7. The phase shifting device of claim 5 in which each reactive termination comprises a grounded series combination of an abrupt junction varactor and an inductor.
8. The phase shifting device of claim 5 in which each reactive termination comprises a grounded series combination of a hyperabrupt junction varactor and an inductor.
9. The phase shifting device of claim 3 in which the frequency multiplier comprises a field effect transistor.
10. The phase shifting device of claim 3 further comprising a second frequency multiplier operatively connected to the first frequency multiplier for receiving the second signal for translating the second signal to a third signal having a third frequency, the third frequency being higher than the second frequency.
11. The phase shifting device of claim 10 in which the third frequency is a harmonic of the second frequency.
12. A method of phase shifting a signal, the method comprising the steps of:
phase shifting a first signal at a first frequency;
and translating the first frequency to a second frequency, higher than the first frequency.
phase shifting a first signal at a first frequency;
and translating the first frequency to a second frequency, higher than the first frequency.
13. The method of claim 12 in which the second frequency is a harmonic of the first frequency.
14. The method of claim 13 in which the second frequency is a carrier frequency and the first signal is a subharmonic of the carrier frequency.
15. The method of claim 12 in which the first frequency is translated in a FET, having a gate bias, the first signal has a signal level, and the gate bias and subharmonic input signal level are selected such that the FET has unconditional stability at all subharmonics of the second signal.
16. A 360 degree linear phase shifter, for continuous modulation of the phase of a carrier signal, comprising:
a linear phase shifter comprising a quadrature coupler having an input port, reflective terminations and an output port, the input port being connected to a source of a reference signal at a subharmonic frequency of the carrier frequency, the linear phase shifter being operable over a fraction of the full 360 degree range, the reflective terminations having a reactance controlled by a linear control signal; and a frequency multiplier operatively connected to the output port for translating the subharmonic reference frequency to the carrier frequency and restoring full 360 degree phase shift range.
a linear phase shifter comprising a quadrature coupler having an input port, reflective terminations and an output port, the input port being connected to a source of a reference signal at a subharmonic frequency of the carrier frequency, the linear phase shifter being operable over a fraction of the full 360 degree range, the reflective terminations having a reactance controlled by a linear control signal; and a frequency multiplier operatively connected to the output port for translating the subharmonic reference frequency to the carrier frequency and restoring full 360 degree phase shift range.
17. The linear phase shifter of claim 16 in which the reference signal has a frequency in the range from 1 to 100 GHz.
18. The linear phase shifter of claim 17 in which the reflective terminations comprise reverse biased varactor diodes.
19. The linear phase shifter of claim 17 in which each reactive termination comprises a grounded series combination of an abrupt junction varactor and an inductor.
20. The linear phase shifter of claim 17 in which each reactive termination comprises a grounded series combination of a hyperabrupt junction varactor and an inductor.
21. The linear phase shifter of claim 17 in which the frequency multiplier comprises a field effect transistor.
22. The linear phase shifter of claim 16 in which the frequency multiplier is a multiple stage frequency multiplier.
Priority Applications (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CA002202457A CA2202457A1 (en) | 1997-04-11 | 1997-04-11 | Microwave phase shifter including a reflective phase shift stage and a frequency multiplication stage |
| US08/843,970 US6111477A (en) | 1997-04-11 | 1997-04-17 | Microwave phase shifter including a reflective phase shift stage and a frequency multiplication stage |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| CA002202457A CA2202457A1 (en) | 1997-04-11 | 1997-04-11 | Microwave phase shifter including a reflective phase shift stage and a frequency multiplication stage |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA2202457A1 true CA2202457A1 (en) | 1998-10-11 |
Family
ID=4160399
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA002202457A Abandoned CA2202457A1 (en) | 1997-04-11 | 1997-04-11 | Microwave phase shifter including a reflective phase shift stage and a frequency multiplication stage |
Country Status (2)
| Country | Link |
|---|---|
| US (1) | US6111477A (en) |
| CA (1) | CA2202457A1 (en) |
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| RU2460183C1 (en) * | 2011-02-21 | 2012-08-27 | Федеральное государственное унитарное предприятие "Научно-производственное предприятие "Исток" (ФГУП НПП "Исток") | Microwave phase changer |
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| RU2367066C1 (en) * | 2008-08-18 | 2009-09-10 | Федеральное государственное унитарное предприятие "Научно-производственное предприятие "Исток" (ФГУП НПП "Исток") | Microwave phase changer |
| RU2460183C1 (en) * | 2011-02-21 | 2012-08-27 | Федеральное государственное унитарное предприятие "Научно-производственное предприятие "Исток" (ФГУП НПП "Исток") | Microwave phase changer |
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| US6111477A (en) | 2000-08-29 |
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